Ferrite transformers and small- to mid-sized loop antennas

I've been doing experiments with a large wire loop (using ordinary zip cord as used for electrical appliances) on my concrete balcony for receiving and, eventually, QRP transmitting purposes. The wire loop is a vertical rectangle, approximately 4 meters wide by 2 meters high, and fed in a corner. The corner feed is not optimal from a balance perspective, but is the most convenient mechanical location for the feedpoint.

Noisy loop 1: Untuned active loop antenna

At first I tried using the wire as a receive-only, active loop antenna along the lines of M0AYF's untuned active loop antenna. I had previously had good experience with this active loop antenna, using a long, 3cm-wide copper strap formed into a square shape having approximately 0.75 meters per side. However, when I now tried the same loop amplifier with the larger 4m x 2m wire loop antenna, the noise was much higher than I expected.

Noisy loop 2: Tuned, passive loop antenna with large air-core transformer (auxiliary coupling loop)

Next I tried tuning the antenna with a small variable capacitor. To get the signal to the receiver, I used a random-length of 2mm-diameter wire, probably about 1 meter long, formed it into a loop, and clipped it to the main antenna element. The coaxial cable going to the receiver was then connected to the ends of the smaller auxiliary coupling loop. This is a common way to feed small transmitting loop antennas. Though this antenna does not qualify as "small", the principle is the same -- a loosely coupled auxiliary loop serves to transfer energy from and to the main resonant loop.

Compared with the untuned active loop antenna, I expected much quieter performance with the tuned loop. However, performance only marginally improved. The noise level was consistently high across the entire tuning range, though strong signals could be brought up out of the noise by peaking the signal with the variable capacitor. Overall performance was very poor.

I then noticed that simply connecting the auxiliary coupling loop to the receiver, even without the coupling loop near the main resonant loop, resulted in an increase in the noise level of the receiver. Therefore, the auxiliary coupling loop -- all by itself -- was picking up some kind of local noise, through electric or magnetic coupling. The noise is probably ambient electric-field and possibly magnetic-field noise produced by electrical appliances of other nearby residents.

A quiet loop: Tuned, passive loop antenna with small ferrite-core toroidal transformer

I decided to try to prevent this stray coupling into the feed system by using a ferrite toroidal core to form a transformer. I am not an expert on the physical mechanisms of noise ingress. However, I thought that using a ferrite transformer could have two potential benefits: first, a smaller physical area occupied by the transformer (compared to the air-cored auxiliary coupling loop transformer), that hopefully would reduced the induced noise; and second, the self-shielding nature of toroidal transformers that keeps the magnetic flux mainly confined to within the core.

I wound the resonant loop's wire element twice through the ferrite core, and made a secondary winding of three turns, that then connected to the coaxial cable leading to the receiver.  For mechanical convenience, the ferrite transformer was located next to the corner-mounted tuning capacitor; such a location for the transformer increases (compared to the typical transformer location diametrically opposite the capacitor) the impedance seen looking into the transformer. Considering both the higher-impedance location of the transformer, and the ad-hoc turns ratio on the transformer, it is certain that the resulting impedance transformation is incorrect (i.e. it does not match exactly to 50 ohms) and will need to be corrected when I adapt the antenna for transmitting. But for receiving, we can tolerate some degree of impedance mismatch.

The result of using the transformer, for receiving, was as hoped: a dramatic decrease in noise levels, along with high signal levels when the capacitor was peaked to the reception frequency. Though the loop balance is compromised by the asymmetrical construction (due to the corner-mounted capacitor, and the capacitor-sited transformer), the loop was nevertheless comparatively quite immune to the local noise, meaning that there was not significant noise ingress via common-mode current.

Therefore, in an environment with high levels of nearby electromagnetic noise, the ferrite transformer method for coupling to a loop antenna may be more immune against near-field noise than a larger, air-core transformer formed with the traditional auxiliary loop.

Analysis: Balance, common-mode currents, chokes

For the other two antenna configurations that were comparatively noisy -- the untuned active loop antenna configuration, or the passive tuned loop antenna with air-core auxiliary coupling loop --  if we assume the cause of the noise is imbalance in the loop, and that the imbalance is allowing the noise to induce common-mode currents, then it is possible that using a common-mode choke on the feedline might be able to reduce the noise to acceptable levels. Due to the geometry of the balcony, perfect symmetry in the loop construction is impossible, and even if it were possible to construct the loop with maximum physical symmetry, the environmental asymmetry will always unbalance the antenna to some extent.

Regarding the auxiliary coupling loop: perhaps one intuitive explanation would be to say that as the auxiliary loop becomes large (due to the main resonant loop also being large), then the auxiliary coupling loop itself starts to become more sensitive to balance, and any imbalance in the construction of or environment around the coupling loop itself will allow common-mode currents to flow. There may also be other mechanisms at play that could explain the noise ingress. There is a construction method that involves shielding the coupling loop; this technique may improve balance within the coupling loop and may reduce the common-mode currents and hence the noise ingress. An experiment, comparing the noise ingress in an isolated coupling loop and in an isolated and shielded coupling loop, could determine if shielding the coupling loop would help. But in this particular case, with this large 4m x 2m wire antenna, there seems to be no practical advantage of continuing to investigate the use of a comparatively large auxiliary coupling loop; a ferrite transformer is more convenient and seems to have no disadvantages.

Varactor-tuned shortwave superhet design, part 1

He who knows, does not speak. He who speaks, does not know. -- Lao Tzu

For several years now I have been meaning to design and build a shortwave superheterodyne receiver that covers 3-30 MHz. I have started and stopped the design process several times, every time hitting up against some problem that seemed too difficult or too tedious to solve. Along the way, I have built a few prototypes, which ended up with unsatisfying performance.

In performing online research for this topic, I have not found much online material that covers the design process for a superheterodyne receiver that fulfills my specifications. My specifications are rather unique (more on that in a moment), and given my chosen specifications, there are a number of rather tricky issues in making a good implementation. 

However, a recent posting on TheRadioBoard (Reference 1) showcased one individual's homebrew shortwave superheterodyne receiver. That posting includes a link to a YouTube video of the set in operation. The set is obviously a good performer, and it is designed with some similar goals as my envisioned design. That inspired me to again resume my own design work.

Reading through that author's description of his set, it is clear that there are a number of subtle issues (like the alignment of the double-tuned front-end RF filter) that need require some experience and intuition to solve. That author obviously has the knowledge to design, build, and align a shortwave superhet. However, that article does not go step-by-step into the construction and alignment process, and also does not explain the design process for the receiver. Hence, I chose the Lao Tzu quote to open this section. Those who know how to design a shortwave superhet don't tend to describe the design process from conception to completion, as I imagine the required skills are already second-nature to such experienced individuals. They just do it. On the other hand, those -- such as myself -- who don't thoroughly know the design process must ask questions, make tentative statements, and seek feedback from others -- to eventually become one who knows, and needs no longer to speak.

Enough philosophy -- on with the design.

My requirements for a shortwave superheterodyne receiver

1. It must cover 3-30 MHz.
2. It must use plug-in coils for bandswitching.
3. It must use toroidal-core inductors for the RF and LO coils, to enable a compact build. Air-core solenoidal coils would require much more physical space.
4. It must use a first intermediate frequency of 2 MHz. The current design will be single-conversion, but a future design may be double-conversion. The chosen IF places the image signal 4 MHz away from the desired signal.
5. It must use a double-tuned front-end RF filter for good image rejection, even at the upper-end of the tuning range (30 MHz). At 30 MHz, an image signal that is only 4 MHz away requires at least a double-tuned filter for acceptable (~50 dB) attenuation of the image signal.
6. It must use varactors (variable capacitance diodes) for tuning of both the front-end RF filter and the local oscillator. This allows simpler mechanical construction than using large air variable capacitors, and opens up future possibilities for remote control of the receiver via control voltages.
7. It must implement proper tracking between the front-end RF filter and the local oscillator. It must not be necessary to separately tune the RF and LO stages; single-knob tuning is required.
8. It must be able to be aligned with no test equipment other than a general-coverage receiver. In particular, no oscilloscope or spectrum analyser is available.

The difficulties posed by my requirements

At first glance, these requirements don't seem so daunting, but once the actual work begins several difficulties arise.

Problem: Adjustment of the toroidal inductors

Traditional superheterodyne receivers (that have a low intermediate frequency) must solve the tracking problem, where a tuned front end always has a peak frequency response that is at a fixed frequency offset from the local oscillator. The fixed frequency offset is, of course, equal to the intermediate frequency of the receiver. To achieve this tracking, typically it is required to adjust not only the tank capacitance but also the tank inductance. Commercial superheterodyne receivers tended to use slug-tuned inductors that could be tuned precisely. However, my set will be using toroidal inductors, that do not allow easy adjustment like slug-tuned inductors. However, even when using toroidal inductors, some degree of adjustment of the tank inductance can be done by adjusting the spacing between turns. Hopefully, this small degree of adjustment will be enough for the RF-LO alignment.

Solution

Stretch or squeeze turns on the toroid to make minor adjustments in the inductance as needed for tracking.

Problem: aligning the two front-end RF tanks without test equipment

To achieve good image rejection, we have not one, but two front-end RF tanks. These both must be aligned with each other, and furthermore they must be aligned with (and at a fixed 2 MHz offset from) the local oscillator. 

Without test equipment like a spectrum analyser, it is difficult to know the exact filter response of the double-tuned RF filter. In particular, we must be very careful not to over-couple the two RF tanks. If we do, we will have a double-humped response that allows unwanted signals to pass through the filter, and that makes filter alignment difficult and confusing (Reference 2).

Therefore we need some way of ensuring the tanks are not over-coupled, and also of ensuring the tanks are in alignment -- without the use of a spectrum analyser.

I ran several LTspice simulations to determine a feasible filter design that could be aligned step-by-step. A future article will cover this topic. For now, I shall summarize my results as follows.

Solution

Use bottom-coupled, capacitive coupling of the tanks to avoid over-coupling and avoid any chance of a double-humped response. For alignment, excite the filter with a broadband noise source and find the peak noise response on a monitoring receiver. Tweak the L and/or C values of the two RF tanks for maximum noise response at the highest frequency tunable by the filter. Because we know the tanks are not over-coupled and will not exhibit a double-humped response, we can be sure that the filter's noise peak occurs only at one frequency and not two frequencies. Then, this peaked response at the highest frequency ensures the best filter alignment between the two RF tanks at the highest filter frequency (where it is most critical for good image rejection), with possible misalignment at the lower filter frequencies (where misalignment is less critical). Finally, align the LO with the RF front-end using a trimmer capacitor, padder capacitor, and inductance adjustment of the LO inductor.

Problem: How to physically switch 3 coils

Using plug-in coils for bandswitching requires switching not one, but three coils -- two front-end coils, and one local oscillator coil. Physically, how should this be accomplished? Each coil could be plugged-in separately, but it would be more convenient to make a combined coil assembly containing all 3 coils required to cover a single band. But, a combined coil assembly with three coils requires a physical connector that has enough pins to switch all three coils.

Solution

Use an 8-pin IC socket for bandswitching. With careful design of the local oscillator, 8 pins are just enough to switch all 3 required coils.

Even when using an 8-pin IC socket, there are physical issues of how to layout the 3-coil assembly to ensure minimum length of the coil leads, which is necessary to minimise stray couplings and ensure good HF performance of the RF filter and the local oscillator. The coil layout should also be physically stable while still allowing access to and adjustment of the coils for tank alignment. A future article will cover this topic.

Problem: Finding high-Q, wide-range varactors for the front-end RF filter

A general observation about varactors is that wide-tuning-range varactors tend to have lower Q. For example, I have used a 1SV149 varactor, which can vary from about 35 pF to 500 pF, in a widely-tuning regenerative receiver that covered several MHz. However, for RF filter use, this varactor has too low Q and will degrade the filter response. We want a varactor that has a Q of at least several hundred at 30 MHz. Most toroidal-core inductors at HF will have Q of around 100-200 (Reference 3, Reference 4, and Reference 5), and we want the varactor Q to be higher than this -- ideally, much higher -- to avoid loading down the resonant tank. Excessive loading of the resonant tanks in the filter would flatten the peak response of the filter and reduce the desired signal's strength in comparison to the undesired image signal's strength, thereby effectively reducing the image rejection of the receiver.

Varactors designed for VHF/UHF use have acceptable Q values at HF, as a perusal of their datasheets will reveal. They however have the disadvantage that their capacitance values are much smaller. The maximum-to-minimum capacitance ratio is also smaller, meaning that VHF/UHF varactors have a smaller tuning range than lower-frequency varactors like the 1SV149.

If the varactor datasheet specifies the varactor Q at VHF, we can estimate with a simple formula the varactor Q at any other frequency (reference: Reference 6, p. 5), and thereby estimate if the Q is high enough for HF filter use. A future article will cover this topic. For now, my results are summarised as follows.

Solution

Use the VHF varactor FV1043 (Reference 7), which has a capacitance range of 10-20 pF and Q of 100 at 100 MHz, giving an extrapolated Q of 333 at 30 MHz, which roughly agrees with the datasheet-given Q value of approximately 650 at 30 MHz. For tuning each of the 3 tanks (2 RF tanks and 1 LO tank), use 5 such varactors in parallel (15 varactors required in total) for a capacitance swing of 50-100 pF for each tank. These capacitance values are reasonable for HF use and allow smaller inductors (than would otherwise be needed with only a 10-20 pF capacitance swing of only one varactor diode) to be used in the RF and LO tanks, requiring less wire and reducing the burden of winding these inductors.

The resulting tuning range of 50-100 pF is not a very wide tuning range, but for HF use it is barely acceptable as it will allow the tank to be tuned over a span ranging from about 1 MHz (at a frequency of 3 MHz) to several MHz (at the highest frequency of 30 MHz). With the 50-100 pF capacitance swing for one band of the receiver, some calculations (to be covered in a future article) reveal that the receiver's entire tuning range of 3-30 MHz can be covered in 8 frequency bands, which is somewhat large but still a reasonable number.

Though I have decided on using FV1043 varactors for this project, ZL2PD has some interesting results on using Zener diodes as varactors at HF (Reference 8).

Problem: Preventing excessive tank voltage in the local oscillator

A significant difficulty with varactor-tuned oscillators is that the varactor itself, providing the tank's resonating capacitance, is a voltage-controlled device. The control voltage is supplied externally by a separate bias voltage to tune the tank. But the oscillating signal voltage in the tank due to can also, to a degree, influence the voltage seen by the varactor and hence influences the tank frequency as well. If the tank voltage is too large, the tank's own signal voltage can significantly affect the varactor's bias and hence lead to waveform distortion. As a rule of thumb, the RF signal voltage should be kept less than 15% of the varactor's bias voltage (Reference 9, p. 10).

An often-used solution to this problem is to use an automatic gain control (AGC) circuit after the oscillator, to keep the oscillator's tank voltages in check. However, the design of AGC control loops is tricky and if we are not careful, the AGC will not operate properly and may become unstable, resulting in periodic and interrupted bursts of oscillator activity (squegging) as the AGC repeatedly tries and fails to regulate the quickly-changing oscillator amplitude. Another difficulty is that the envisioned receiver should work over 3-30 MHz, which means that an AGC control loop would need to be designed that works properly over this entire range with all of the different coils that are used to cover this wide frequency range. I expect this is not an easy task to design a stable AGC that works over this entire range with a variety of coils. I consider it too risky to attempt oscillator AGC in the first receiver design. I may attempt this in a future receiver design, however.

Another solution is to use a hybrid-feedback oscillator design that passively tries to equalise the amount of oscillator feedback across the entire tuning range (Reference 10, p. 15). Unfortunately, this approach is also difficult. (Note added 2016-09-04: but I believe I found a solution; see section "Alternative solution" below.) First, there is the above-mentioned difficulty of supporting a wide frequency range with several coils. Second, the hybrid feedback approach, when applied to an LC tank that is tracked as part of a superheterodyne receiver, has an inherent limitation that one of the capacitors must serve a dual role both as the padder capacitor (to limit the capacitance swing of the tuning capacitance) and as the Vackar feedback capacitor (providing Vackar-oscillator-style capacitive feedback in addition to the tickler feedback). This dual-use makes it difficult to select an appropriate value for the capacitor that both provides the proper amount of feedback (not too much or too little) and properly limits the capacitance swing as required for RF-LO tracking. 

Finally, another solution -- also mentioned Reference 10 -- is simply to damp the tank. This is not ideal, as it reduces the Q of the tank and hence compromises the oscillator's signal purity. But it is a simple solution, it can be applied easily to any LC tank at any frequency, and LTspice simulations show that it is able to keep the tank oscillation amplitude mostly low and mostly constant over the range of the oscillator. A future article will cover this topic.

Solution

Use an Armstrong oscillator, and damp the tank with a low-value parallel resistance. Decrease the resistance (increasing the damping) until, at the lowest oscillator frequency, the oscillator just barely starts. This ensures a very small oscillator amplitude (ideally less than 15% of the varactor bias voltage) at the lowest frequency, where varactor bias will be the lowest (around 1 volt). As the varactor bias increases, tuning the oscillator upwards in frequency, the Armstrong oscillator will show an increased oscillation amplitude, though the damping resistor will largely keep this in check, ensuring the amplitude only grows slightly. Furthermore, even if the amplitude does grow slightly with increasing frequency, the varactor bias is also greater at higher frequencies, and therefore the varactor is more resistant to detuning by the oscillating signal voltage.

Alternative solution (2016-09-04)

After extensive LTspice simulation I believe I have found a better solution than tank damping for keeping the oscillator's tank voltages low. The solution outline is as follows.

1.  Use a BJT oscillator instead of a JFET oscillator. A JFET may have too little gain in the desired oscillator configuration.
2. Configure the oscillator to be a Seiler-Vackar hybrid feedback oscillator (Reference 10, p. 16). Do not attempt to re-use the padder capacitor as part of the feedback network; use it only for tracking purposes (to limit the oscillator tuning range).
   
   As mentioned above, circuit simulations indicated that in this oscillator configuration, the JFET (using a 2N4416 in the simulation) did not always have sufficient gain for oscillation at some frequencies, but a BJT did.
   
   The reason for the Seiler-Vackar hybrid instead of the Armstrong-Vackar hybrid is that Armstrong feedback requires a tickler coil, and the number of turns on the tickler coil cannot be reduced to less than one. This might lead to an excess of Armstrong-style feedback, especially at higher frequencies, and when using a tickler it would be impossible to reduce this feedback below one turn (especially when using a toroidal-core inductor that has a high coupling coefficient). On the other hand, using capacitive feedback only as in the Seiler-Vackar allows both the Seiler (Colpitts-style) feedback path and the Vackar feedback path to be adjusted finely, by altering the capacitance values. Trimmer capacitors or gimmick capacitors can be used for very fine adjustments of the feedback.
3. Control base bias with a potentiometer as a temporary gain control (similar to the regeneration control on a regenerative receiver), to gauge the oscillation threshold as the LC tank is tuned across its range.
4. Using the base bias potentiometer to gauge the oscillation threshold, tune the LC tank across its tuning range and balance the Seiler feedback path and the Vackar feedback path such that a uniform, weak oscillation amplitude is achieved across the entire tuning range.
5. Finally, the capacitive divider C3/C4 (Reference 10, p. 16) can be used to tweak the oscillator's loop gain (independently of the transistor's base bias) so that oscillation just barely commences. Because this capacitive divider will be part of the plug-in coil assembly, this allows the loop gain to be tweaked individually for each set of plug-in coils (independently of the transistor's base bias), thereby ensuring that every set of plug-in coils will exhibit uniform, weak oscillation across its entire tuning range. Tweaking the feedback with the C3/C4 divider therefore allows setting the transistor bias to a fixed value that need not be adjusted when the plug-in coils are changed to switch bands.

Other issues to consider

In addition to the above problems, the following are additional design issues that must be decided.

1. Choice of mixer. A mixer that is less prone to distortion (such as a level-7 diode ring mixer) will require higher output power from the local oscillator.
2. Ensuring adequate LO power for the mixer. This may require amplifiers and buffers after the LO, especially considering that we are intentionally keeping the LO tank voltages low to avoid distortion from the varactor.
3. IF filtering. Following a strong mixer with a broadband amplifier and a crystal filter will allow good basic receiver performance and reception of weak signals near strong signals (Reference 11). A future version of this receiver may investigate use of a single-crystal filter. A multiple-crystal filter would allow better filter response and broader bandwidth (for less critical tuning), but requires precise measurements of crystal parameters which may be difficult with my limited test equipment.
4. Detection. The first version of this receiver will use a 2 MHz regenerative detector. This will serve both as an IF filter and as a detector. It may be necessary to precede this stage with a buffer amplifier. For best performance, that buffer amplifier may also need to be tuned.
5. AGC. The first verison of the receiver will likely not use AGC, but a future version may incorporate AGC, by reducing the gain or attenuating the input of either the IF or the RF stage.

Next steps

This was a long article -- and this only covers the basics of the design.

Future articles will go into more detail about each stage's design, analysis, construction, and alignment. I plan to start with the double-tuned RF filter.

References

1. User "sixtynine." Discussion topic titled "superhet regen." TheRadioBoard forums. http://theradioboard.com/rb/viewtopic.php?f=3&t=7163. August 26, 2016.
2. Hayward, W. "The double-tuned Circuit: An experimenter's tutorial." http://www.robkalmeijer.nl/techniek/electronica/radiotechniek/hambladen/qst/1991/12/page29/index.html. December, 1991.
3. Cox, J. "Iron Powder Cores for High Q Inductors." http://www.micrometals.com/appnotes/appnotedownloads/ipc4hqi.pdf. Undated.
4. Amidon Associates, Inc. "Iron-powder toroidal cores Q-curves." http://www.amidoncorp.com/product_images/specifications/1-18.pdf. Undated.
5. Amidon, Inc. "Iron powder toroidal curves, Typical 'Q' curves." http://www.amidoncorp.com/product_images/specifications/1-11.pdf
6. Skyworks Solutions, Inc. "Application Note, Varactor Diodes." http://www.skyworksinc.com/uploads/documents/200824A.pdf. August 15, 2008.
7. Semtech Electronics Ltd. "FV 1043 Tuner AFC Diode." http://radio-hobby.org/uploads/datasheet/659/fv10/fv1043.pdf. Undated.
8. Woodfield, A. "ZL2PD Hunts for Varicap Diodes." http://www.zl2pd.com/Varicaps.html. January/February 2007.
9. Hollos, S., and Hollos, R. "Using varactors." http://www.exstrom.com/journal/varac/varac.pdf. 2001.
10. N., Vlad. "Frequency compensated LC networks for oscillators with the wide tuning range." http://www.kearman.com/vladn/hybrid_feedback.pdf. February 1, 2012.
11. Newkirk, D. Discussion topic titled "OOPS!" Regenrx forums. https://beta.groups.yahoo.com/neo/groups/regenrx/conversations/messages/23026. March 1, 2015.